Low loss current sensor and power converter using the same

ABSTRACT

A low-loss current sensor for use with a circuit containing a capacitor to sense a current flowing into a node in the circuit is provided. The current sensor includes a differentiator circuit having an input connected to the circuit capacitor and adapted to generate an output which is proportional to the current flowing through the circuit capacitor. The novel use of a capacitive current divider allows the sensor to sense current with virtually no power dissipation.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority under 35 U.S.C. §119 to U.S.provisional patent application Ser. No. 62/200,194 on Aug. 3, 2015,which is incorporated by reference herein in its entirety.

TECHNICAL FIELD

The present invention relates to current sensors for electrical circuitscontaining a capacitor, and in particular current sensors for use withpower converters.

BACKGROUND OF THE INVENTION

In electrical power converters, the measurement of current is commonlyachieved by the use of a low-value shunt resistor to generate a smallvoltage proportional to the current flowing through the shunt resistor.This voltage is then generally amplified for further use or measurement.In a power converter, such a current sensor can be used to measure thecurrent flowing through an inductor, which is used as a magnetic energystorage or transmission element.

However, one problem of using a conventional current sensor is that itwastes power, which may be significant in high-efficiency converters.Therefore, it would be desirable to provide a device and method for moreefficiently sensing the current of an electrical circuit.

SUMMARY OF THE DISCLOSURE

According to one aspect of the present invention, a low-loss currentsensor for use with a circuit containing a capacitor to sense a currentflowing into a node in the circuit is disclosed. The current sensorincludes a differentiator circuit having an input connected to thecircuit capacitor and adapted to generate an output signal which isproportional to the current flowing into the circuit capacitor. Thenovel use of a capacitive current divider allows the sensor to sensecurrent with virtually zero dissipated energy.

According to another aspect of the present invention, a power converter(e.g., DC-DC converter) utilizing the above low-loss current sensor isprovided. The power converter has two serially connected switchesdefining a common node therebetween and first and second capacitorsconnected across respective first and second switches. The capacitorscan be separately connected components or are part of the switches suchas the small signal output capacitance inherent in the switches. Aninductor is connected to the common node and is adapted to drive a loadat an output such as a battery.

The current sensor includes a sensing capacitor and resistor. Thesensing capacitor has a first terminal connected to the common node andis adapted to generate an output carrying a current which isproportional to the current flowing into the first and secondcapacitors. The resistor is connected to the second terminal of thesensing capacitor and defines an output of the current sensor, a voltagedrop across the resistor being substantially proportional to the currentflowing through the sensing capacitor. A controller switches the firstand second switches in a complementary manner and controls the turn-ontime of at least one of the first and second switches based on readingsof the current sensor output.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram of a zero voltage switching powerconverter with a controller according to an aspect of the presentinvention.

FIG. 2 illustrates switching waveforms for a conventional hard-switchedpower converter.

FIG. 3 illustrates switching waveforms for a ZVS power converteraccording to an aspect of the present invention.

FIG. 4 illustrates switching waveforms for an optimized ZVS powerconverter according to an aspect of the present invention.

FIG. 5 illustrates dead times and body diode conduction periods for aZVS power converter according to an aspect of the present invention.

FIG. 6 is a functional block diagram of a controller for controlling apower converter according to an aspect of the present invention.

FIG. 7 is a functional block diagram of a low loss current sensoraccording to an aspect of the present invention.

FIG. 8 illustrates an exemplary low loss current sensor according to anaspect of the present invention.

FIG. 9 illustrates representative waveforms of various nodes in thecurrent sensor of FIG. 8.

DETAILED DESCRIPTION OF THE INVENTION

Definitions

Switch: While the specification illustrates a metal-oxide-semiconductorfield-effect transistor (MOSFET) as an exemplary switch, a switch canrefer to any electrical element capable of conducting or blocking acurrent, such as a bipolar transistor, IGBT, relay, diode, and the like.

Switching node: An electrical node in the switching system at thejunction of two or more switches and an inductive element or otherelectrical load or source.

Inductor: An explicit inductor, or the intrinsic inductance of anotherelectrical component or circuit.

Transition time: The time during which the switching node is in betweenthe switching potentials, determined by the inductor current, switchcurrent(s), capacitance at the switching node, and the differencebetween the applicable switching potentials.

Switching potentials: The potentials between which the switching node isswitched by the switches.

Controller: a processor, microcontroller or other digital or analogcontrol circuit capable of controlling the switches based on the variouscontrol inputs and the desired goals of system operation.

Body diode: The body diode of a MOSFET, or, for other switch types, animplicit or explicit diode or other device or mechanism that allowscurrent to flow through the switch independent of the switch controlsignal.

In the interest of increasing the efficiency and decreasing the size ofpower converters and other electrical devices which switch an inductorbetween different voltages, the technique of Zero Voltage Switching(ZVS), one of the techniques commonly referred to as Soft Switching, hasbeen developed. ZVS is a technique that reduces switching loss, thecomponent of system loss that arises during the time period (called thetransition time) while the inductor is switched from one potential toanother and during which a switch may have both a substantial voltageacross it and a substantial current through it, resulting in substantialpower dissipation.

FIG. 1 shows a typical switching circuit 20 capable of ZVS operation. C1and C2 are the implicit or explicit capacitances (either of which may bezero) in parallel with switches S1 and S2. The switches S1 and S2 can beMOSFET switches and are connected in series, with a switching node Ndefining a common node. The switches S1 and S2 are connected to and canbe driven by a controller 40, which typically provides complementarysignals such that only one switch is closed at any one time. If S1 andS2 are MOSFETs, then their gates will be connected to the controller 40.

In the embodiment shown in FIG. 1, inductor L is switched between V1 andground (0V) to transfer power to or from another potential V2. Forexample, V1 could be a solar panel that generates power and V2 could bea battery to be charged from the solar panel.

In many conventional hard-switched switching circuits, the currentthrough the inductor L is substantially constant (i.e., the AC componentof the current is significantly smaller than the DC component). FIG. 2illustrates switching waveforms for a typical hard-switched powerconverter. Waveform 56 shows a voltage level at node N while waveform 58shows current through the inductor L.

Waveform 56 shows that the converter 20 (if it were driven as ahard-switched system) switches the switching node N between 20V andground (0V) while waveform 58 shows the DC output current at 5 A and ACcomponent at 3 App (−1.5 A to +1.5 A from the DC level), resulting in aminimum inductor current Ilmin of 3.5 A. Since the AC p-p component isless than twice the DC component, the current does not reversedirection, and the minimum current is positive.

By contrast, in a ZVS circuit, the amplitude of the AC component of thecurrent through the inductive component is more than twice the magnitudeof the DC component, which means the current reverses direction for atleast a portion of each cycle of the switching converter 20.

A second defining feature of a ZVS power converter is the use ofcapacitance in parallel with one or more of the switching elements. Thiscapacitance may take the form of an added capacitor, or may be intrinsicto the device, such as the output capacitance (Coss) of a MOSFET.

The reversing current and the capacitance across one or more of theswitches allows the ZVS switching circuit to take on some of theproperties of a resonant L-C circuit. Rather than relying on theswitches S1 and S2 to pull the switching node N from one potential tothe other, the power converter 20 employs the reversing current in theinductor L. When each switch turns off, the current from the inductor Lflows through the capacitance (C1 and C2). This capacitance slows thevoltage slew rate of the switching node N, allowing the switch to turnoff before the voltage across it has increased appreciably, therebysubstantially eliminating switching loss. Similarly, each switch iscontrolled by the controller 40 to turn on only when the voltage acrossit (e.g., voltage between drain and source of a MOSFET switch S1 or S2)is approximately zero, again substantially eliminating switching loss.

FIG. 3 illustrates switching waveforms for a ZVS power converter 20according to an aspect of the present invention. The switchingpotentials, DC current, and frequency are identical to those in FIG. 2.However, the p-p AC current as shown in waveform 60 is much larger at 22App, resulting in a minimum inductor current Ilmin of −6 A. Because thecurrent through inductor L becomes negative, the current may be used todrive node N from 0V to 20V while the switches are not conducting, andZVS is thus possible. Conceptually, the converter of the previousexample can be changed to the operation of FIG. 3 by reducing theinductance of the inductor L by a factor of 22/3. In the example shownin FIG. 3, the AC current amplitude is much larger than twice the DCcomponent, resulting in unnecessary loss.

FIG. 4 illustrates switching waveforms for an optimized ZVS powerconverter 20 according to an aspect of the present invention. Waveform62 shows the voltage level at node N while waveform 64 shows the currentthrough the inductor L. As can be seen by waveform 62, the powerconverter 20 is operated at twice the frequency of FIG. 3. Now, the ACcomponent of the inductor current is 11 App, which is just slightlylarger than twice the DC component. At this operating point, the minimuminductor current Ilmin becomes only slightly negative. Two qualitativedifferences are noticeable in the waveforms 62 and 64: (1) the risingedge of the switching node N voltage in waveform 62 becomes visiblysloped due to the relatively small inductor current driving thetransition, and (2) in turn the bottom corner of the inductor currentwaveform 64 becomes noticeably rounded due to the slow voltagetransition. This is an optimal operating point for a ZVS converter, asthe AC inductor current is just above the threshold necessary for ZVSoperation, and the associated AC losses are at approximately the minimumpossible.

Conventionally, ZVS converters are operated at a fixed frequency. Givenfixed input and output voltages and fixed component values, the ACcurrent flowing through the inductive component is essentially constant,independent of the DC current. An exemplary ZVS converter is disclosedin an article entitled “A Zero-Voltage-Switching Bidirectional BatteryCharger/Discharger for the NASA EOS Satellite” by Dan M. Sable, Fred C.Lee and BO H. Cho in pages 614-621, Applied Power Electronics Conferenceand Exposition, Conference Proceedings 1992, which is incorporatedherein by reference.

At light loads, the fixed losses caused by the AC currents will resultin poor system efficiency as discussed in the above article.Accordingly, techniques are sought that will preserve ZVS operationwhile minimizing the AC currents in the ZVS system.

In most practical applications for ZVS systems and power converters ingeneral, the operating potentials and currents are determined byexternal factors. The input and output potentials will determine the ZVSconverter duty cycle to within a narrow range. However, operatingfrequency remains an independent and free variable.

In one aspect of the present invention, the controller 40 dynamicallyvaries the operating frequency of the switches S1 and S2 on a continuousbasis such that the peak-to-peak inductor current Ilpp is regulated tojust somewhat more than twice the DC inductor current Ildc in order tokeep Ilmin just slightly negative, based on input parameters asdiscussed below. The AC and DC inductor currents may be measureddirectly or calculated from other system parameters.

In one embodiment, the DC current through the inductor is measured, andthe AC current is calculated according to conventional means. Taking thepower converter 20 in FIG. 1 as a representative example, the optimalfrequency f can be calculated as follows.

Assume the transition time is negligible, and S1 and S2 are controlledoppositely with respective on times T1 and T2. These variables arerelated to the frequency f and input and output voltages as follows:

f=1/(T1+T2)   (1)

V2/V1=T1/(T1+T2)   (2)

From basic power electronics:

Ilpp=T2*V2/L, and therefore   (3)

T2=Ilpp*L/V2   (4)

According to our prior definitions:

Ilmin=Ildc−Ilpp/2   (5)

To achieve ZVS, Ilmin<0. Ilmin may be set to a fixed target valueIlmintarget, which may typically have a magnitude in the range of 1% to10% of Ilpp. However, Ilmintarget may depend on the specifics of theconverter, and may also be dynamically optimized for variations in Ildc,V1, V2 and/or other system parameters for additional benefit.

After substitutions and eliminations, the following equation (6) for T2can be obtained:

T2=2*L*(Ildc−Ilmintarget)/V2   (6)

In that case, the following equation, derived from (2), may be used todetermine the on time T1 of switch S1 according to the actual or desiredinput and output voltages V1 and V2:

T1=V2*T2/(V1−V2)   (7)

Thus, the optimal switch timings T1 and T2 may be calculated by thecontroller 40 from the following measured, desired, or constantparameters: V1, V2, Ildc, Ilmintarget, and L.

As can be appreciated by persons of ordinary skill in the art, the powerconverter 20 of FIG. 1 with frequency optimization can provide highefficiency across different load conditions. For example, a typicalfixed frequency ZVS converter, as exemplified in the preceding Sablereference, may have peak efficiencies in the high nineties. However, at10% of full load, electrical efficiency decreases to only about 83%. Bycontrast, the power converter 20 of FIG. 1 with variable frequencyswitching may also have peak efficiencies in the high nineties, but at10% of full load, the efficiency may remain as high as 98%. Thisrepresents almost a tenfold reduction in wasted power at the 10% loadcondition.

Many inductors vary in inductance with current and temperature. Inanother alternative embodiment of the invention, the control scheme isgeneralized as follows in which L is a function of Ildc, Ilpp, and ameasured inductor temperature t1:

T2=2*L(Ildc, Ilpp, t1)*(Ildc−Ilmintarget)/V2   (8)

The inductor currents Ildc and Ilpp can be calculated as above, ormeasured in a conventional manner by a current sensor 66 connected inline with the inductor L and connected to the controller 40 while theinductor temperature can be measured with a temperature sensor 67thermally coupled to the inductor L and electrically connected to thecontroller (see FIG. 1).

In this embodiment, rather than determining T2 with L as a constant, thecontroller estimates L based on current and temperature according to apre-determined relationship.

In the description above, the transition time was taken to benegligible, and switches S1 and S2 were assumed to be controlledoppositely (i.e., as complements of each other). This is a goodapproximation for the overall control methods described above. However,real switches typically have a non-zero switching time between their onand off states. To avoid both switches being on at the same time withdeleterious consequences, practical ZVS and hard switching systemstypically include a dead time, during which both switches S1 and S2 areoff. These dead times are typically small compared to the overallswitching period. In ZVS systems, the dead time allows the inductorcurrent L to drive the switching node N through the switching transitionwithout interference from the switches S1 and S2. Dead times aretypically implemented as delays to the turn on of the respectiveswitches and are typically set to constant values with a large enoughmargin to ensure that the switches don't conduct simultaneously, takinginto account differing loads, temperatures, production variations, andthe like.

In the case where the switches are MOSFETs (or otherwise include anantiparallel diode), after the switching transition, but before theswitch turns on, the MOSFET body diode conducts, resulting in wastedenergy. In systems using other switches, inappropriate dead times canresult in switches closing with a large voltage across them, alsoresulting in wasted energy.

FIG. 5 illustrates typical dead times (rising edge dead time 66 andfalling edge dead time 68) and body diode conduction periods (70 and 72)for a ZVS power converter according to an aspect of the presentinvention.

In one embodiment, the present invention includes a control scheme thatdynamically optimizes dead times to substantially eliminate dead timeerror. Accordingly, body-diode conduction or other similar losses aresubstantially eliminated.

In one embodiment, dead times are calculated based on inductor currentsduring the respective switching transitions, which may in turn becalculated based on measured DC current, switch timings, and input andoutput voltages, according to conventional techniques, one of which isillustrated below.

The definition of a capacitor gives us:

I=C*dV/dt   (9)

Integrating and rearranging, the following formula can be obtained:

Δt=C*ΔV/I   (10)

Δt is the time the switching node N takes to move from ground to V1 orvice versa. Thus, for an ideal dead time DT to minimize body diodeconduction, ΔV is simply V1, I is the average inductor current duringthe transition, and C is the total capacitance at node N, which in thecase of the power converter 20 equals C1+C2 and any parasiticcapacitances. As a first approximation, it can be assumed that theinductor current doesn't change during the dead times, because they arecomparatively brief. In this example, while voltage at node N is rising,the inductor current is at a minimum, Ilmin, and thus, ignoring signs,the transition time and ideal dead time will be:

DT1=C*V1/Ilmin   (11)

While node N is falling, the inductor current is the DC value plus halfthe peak to peak value, and thus, neglecting signs, the ideal dead timewill be:

DT2=C*V1/(Ildc+Ilpp/2)   (12)

Once the ideal dead time has been determined, the controller 40 canoutput appropriate signals to control the gates of the switches S1 andS2 to the determined dead time periods. The parasitic componentscontributing to C may depend on V1 or other system parameters, and thecontroller 40 may estimate these changes based on a pre-determinedrelationship for additional accuracy.

Dead times calculated in this manner can enable performance increasesrelative to the conventional technique of fixed dead times. However, thetechnique is still prone to production variations and other unknowns,meaning a safety margin, resulting in body diode conduction andassociated loss, will still need to be added in practical converters.For example, a safety margin of 10-100% of the determined optimal deadtime can be added in order to allow for measurement or estimation errorsin the parameters used to calculate the optimum dead times.

One method to change the dead time in both digital and analog systems isto delay the turn on of each switch S1 and S2 by the duration of thedesired dead time for that transition. In digital systems, these delayvalues may typically be set directly according to the previouscalculations, with an added margin if desired. In this manner, T1 and T2are substantially unchanged, so there will be no interference with thefeedback loops controlling them, and thus the dead times may beoptimized independently to maximize converter efficiency.

In power converters using MOSFET switches, the voltage across the switchwhen it is turned on will be typically smaller than 0.7V. When theswitch is off and the body diode is conducting, the voltage on theswitch will generally be 0.7-1.5V. Therefore, it is possible todetermine if the body diode is conducting simply by comparing thevoltage across the MOSFET to a threshold, which may be dynamicallyvaried to suit operating conditions such as current, temperature, andthe like. For the case of an N-channel MOSFET, for instance, the bodydiode is conducting if Vds<−0.5V.

In another embodiment, the invention includes a controller 40 whichdetermines dead times via direct measurement of body-diode conduction,resulting in optimal dead-time values at all operating points and overproduction variations, resulting in robustly efficient operation.

FIG. 6 is a functional diagram of a controller 40 for controlling apower converter 20 according to an aspect of the present invention.

The controller 40 of the present invention is connected to various nodesof the power converter 20 through communication links 52 which areconnected to an I/O interface 42, which receives information from andsends information over the communication links. Inputs which are analogin nature may pass through an A/D converter 16 prior to being fed intothe I/O interface 42.

The controller 40 includes memory storage 44 such as RAM (random accessmemory), processor (CPU) 46, program storage 48 such as Flash, FPGA, ROMor EEPROM, and data storage 50 such as a hard disk, all commonlyconnected to each other through a bus 53. The program storage 48 stores,among others, a power control module 54 containing software to beexecuted by the processor 46. The power control module 54 receivesvarious signals from the power converter 20 and controls the conversionratio V2/V1 of the converter based on those signals as will be discussedlater herein.

The power control module 54 may include a user interface module thatinteracts with a user through the display device 11 and input devicessuch as keyboard 12 and pointing device 14 such as arrow keys, mouse ortouchpad. The user interface module assists the user in programming themodule 54 for desired performance of the power converter 20. Any of thesoftware program modules in the program storage 48 and data from thedata storage 50 can be transferred to the memory 44 as needed and isexecuted by the CPU 46.

One exemplary controller 40 may be the AVR series of microcontrollersfrom Atmel Corporation of San Jose, Calif. However, any suitable digitalsignal processor, processor or microcontroller can be used.

According to an aspect of the present invention, a current sensor 100which uses virtually no energy/power is disclosed. As shown in FIG. 7, adifferentiator circuit 102 has an input connected to node N (see FIG.1). An output 104 of the differentiator 102 creates a signal D which isproportional to the rate of change of voltage at node N. Because theimpedance at node N is capacitive during the converter dead times, thesignal at node N is proportional to the current flowing into node Nduring the dead times. Under the control of the controller 40, thissignal D at output 104 is sampled during the dead times for a measure ofinductor current, or fed to a peak detector 106 so as to measure themaximums or minimums automatically. The sampled signal at node 104 orpeak-detected output signal at output 108 can then be used by thecontroller 40 of the power converter 20.

FIG. 8 illustrates an exemplary low loss current sensor according to anaspect of the present invention. The differentiator 102 has beenapproximated by a high-pass filter composed of sensing capacitor C3(typically 1 pF-1 nF) and resistor R1 (typically 100Ω-10KΩ), and thepeak detector 106 has been implemented with diode D1 whose input isconnected to the output 104, and detection capacitor C4 (typically 100pF-1 nF) and switch S3 (which may be a MOSFET) connected in parallel toeach other. The switch S3 is under the control of, or may be an integralpart of the controller 40.

In operation, prior to sensing a new current value, S3 is closed todischarge detection capacitor C4, then opened by the controller 40. Ifoutput 108 feeds into a microcontroller, it may be possible for themicrocontroller to hold the output node 108 low with an internal switchto discharge detection capacitor C4, thus eliminating the need for aseparate switch S3.

During each converter dead time, the full current of inductor L flowsinto node N, with a total capacitance of typically 1-1000 nF, whichincludes the output capacitance Coss of the power switches S1 and S2 andthe explicit capacitors C1 and C2 (see FIG. 1). Some small fraction ofthe inductor L current also flows through sensing capacitor C3,according to the ratio of C3 to the total capacitance at node N, whichmay typically be in the range of 1/100- 1/10,000.

This current is converted to a voltage by resistor R1, rectified bydiode D1, and over several cycles, charges up detection capacitor C4 toa voltage equal to the peak value of output 104 less the voltage drop ofdiode D1 to produce a signal at output 108, which represents a measureof current flowing into the converter capacitors C1 and C2 through theinductor L during one of the dead times. The dead times typically occurat times of maximum and minimum inductor current. Thus, depending on theorientation of diode D1, the invention may be configured to produce ameasure of either the maximum or minimum (Ilmin) inductor current overthe switching cycle.

As can be appreciated by persons of ordinary skill in the art, a typicalmethod of measuring current involves passing the entire current througha resistor, which can be very lossy. By contrast, in the current sensor100 of the present invention, only about 1/1000th of the current isbeing passed through a resistor for only perhaps 1/100th of the time(the converter dead times). Therefore, the current sensor of FIG. 8allows inductor L current to be sensed with extremely low powerdissipation, e.g., on the order of 1/10,000th- 1/100,000th of theconventional technique.

Although the differentiator circuit 102 has been shown with a high passfilter comprising capacitor C3 and resistor R1, other types of circuitscan be used such as a resistor connected in series with an inductorbetween the switching node N and reference with the node common to theresistor and inductor being its output. Conventional activedifferentiators and peak detectors may also be employed.

FIG. 9 illustrates representative waveforms of various nodes in thecurrent sensor of FIG. 8. As in FIG. 1, as the switch S1 turns on andoff, the voltage 110 at node N alternates between 20 V and 0V. Thecurrent 112 flowing through the inductor L rises to approximately 11 Awhen the switch S1 turns on and falls to approximately 0 A when theswitch S1 turns off and S2 turns on. During the dead time when bothswitches S1 and S2 are off, the current through the inductor L fallsbelow 0V to approximately −1.5 A before rising to 0 A.

During the dead time, the output 104, illustrated in waveform 114, risesto approximately 1V while the output 108, illustrated in waveform 116,rises to approximately 0.8V due to the voltage drop across diode D1.Although FIG. 9 shows a relatively horizontal line 116 for output 108,as it would be in steady state, the voltage does rise gradually throughthe initial dead times after the switch S3 discharges the detectioncapacitor C4.

Although the current sensor 100 can be configured to calculate eithermaximum or minimum current (negative or positive peak current) into acapacitive node by changing, for example, the orientation of the diode,in the preferred embodiment for use in the ZVS converter describedherein the invention provides a measure of the minimum inductor currentIlmin. Ilmin can be calculated from the voltage V108 at output 108 ofthe current sensor 100 using the following equation, where M is aconstant (negative in this example) dependent primarily on the circuitcomponent values, and Vk is a constant voltage substantially equal tothe diode drop of diode D1:

Ilmin=M*(V108+Vk)   (13)

Direct measurement of Ilmin allows the controller 40 to regulate T1 andT2 according to equations (6) and (7) so that Ilmin approaches theoptimal value Ilmintarget.

Using Ilmin and the other known or calculated operating parameters, thecontroller 40 can calculate the actual DC current Ildc, which can beused by the controller 40 to control the power converter. Based onequations (4) and (5), the following equation is derived.

Ildc=Ilmin+T2*V2/L/2, or:   (14)

Ildc=M*(V108+Vk)+T2*V2/L/2   (15)

Assuming output V2 stays relatively constant, DC output currents Ildc atdifferent input voltages V1 may be compared without math simply byadjusting the high switch S1 on time T1 only. Changing T1 does notdirectly affect any of the parameters in equation (15), so a change upor down of V108 corresponds directly to a change down or up of Ildc.

According to another aspect of the invention, the controller 40 uses thesensed current at output 108 to control the power converter 20 foroptimal charging of a battery at V2 from a power source at V1. Since theoutput 108 is an analog signal, it is routed to the A/D converter 16 ofthe controller to convert it to a digital signal. Often, the optimalinput voltage at V1 for charging a battery at V2 changes according tothe power being generated by, or other characteristics of, the powersource at V1. For example, a power source such as a solar panelgenerates maximum power at a particular voltage which depends on variousfactors such as insolation, shading, dirt on the panel, temperature, ageof the panel, and the like. That optimal voltage at any given time needsto be found for maximum transfer of power to the load at V2.

One method for finding the optimal voltage V1 is known as hill climbingor Perturb and Observe (P & O). To implement this method, the controller40 can vary the high switch S1 on-time (T1) by a certain increment inone direction, e.g., gradually increasing T1 by a predeterminedincrement thereby gradually decreasing the voltage V1. At the same time,the controller 40 monitors the output 108 to see if Ildc increasesaccording to equation (15). If so, the controller 40 continues toincrease T1 to reduce the voltage at V1 further to continue to “climbthe hill” and increase Ildc. Assuming the battery voltage V2 isrelatively constant, an increase in Ildc corresponds to an increase inpower delivery from the solar panel to the battery.

When the controller 40 determines via monitoring output 108 that Ildchas decreased, it indicates that the power converter 20 went past theoptimal voltage V1 and started to go “downhill” in power delivery. Inthat case, the controller 40 changes direction and starts to decreasethe T1 by a second predetermined increment thereby gradually increasingthe voltage V1. The second predetermined increment can be a smallerincrement for fine tuning the proper T1. The decrease in T1 continuesuntil power output decreases, at which point the controller will beginincreasing T1, repeating the cycle. As can be seen, hill climbing is aniterative process and the controller 40 is continuously adjusting T1 toadjust for the changing power generation condition of the power source.

During this procedure, it may be necessary for the controller 40 toadjust T2 (and T1 proportionally) to maintain Ilmin near Ilmintarget andthus ensure optimal electrical efficiency of the ZVS converter.

It is significant to note that since the change in voltage output at 108is proportional to the change in Ildc, the controller can directlycompare the present voltage value (sampled values) at 108 to a previousvoltage value at 108 to determine whether the current Ildc went up ordown without any further calculation. In other words, adjustment of T1and T2 for both optimal converter frequency and to implement a hillclimbing algorithm may be accomplished without the need to calculateIldc according to equation (15).

In one embodiment, the controller 40 samples the current from thecurrent sensor 100 whenever the readings are needed. In anotherembodiment, the controller 40 may sample the current from the currentsensor 100 at only the dead times, either 66, 68 (see FIG. 5) or both.

According to another aspect of the invention, the power converter 20 canbe operated in a reverse direction to supply power to V1 from V2 simplyby changing the direction of diode D1 in the peak detector 106 andadjusting the controller algorithm appropriately. Two current sensorsmay be used in order to produce a bidirectional power converter.

The foregoing specific embodiments represent just some of the ways ofpracticing the present invention. Many other embodiments are possiblewithin the spirit of the invention. Accordingly, the scope of theinvention is not limited to the foregoing specification, but instead isgiven by the appended claims along with their full range of equivalents.

What is claimed is:
 1. A low-loss current sensor for use with a circuitcontaining a capacitor to sense a current flowing into a node in thecircuit, comprising: a differentiator circuit having an input connectedto the circuit capacitor and adapted to generate an output which isproportional to the current flowing through the circuit capacitor. 2.The low-loss current sensor of claim 1, wherein the differentiatorcircuit includes a sensing capacitor having a first terminal connectedto the circuit capacitor and a second terminal carrying a current whichis proportional to the current flowing through the circuit capacitor. 3.The low-loss current sensor of claim 1, wherein the differentiatorcircuit includes: a sensing capacitor having a first terminal connectedto the circuit capacitor and a second terminal, the sensing capacitorcarrying a current which is substantially proportional to the currentflowing through the circuit capacitor; a resistor connected to thesecond terminal of the sensing capacitor, a voltage drop across theresistor being proportional to the current flowing through the sensingcapacitor.
 4. The low-loss current sensor of claim 1, further comprisinga peak detector coupled to the differentiator output to detect a peakvalue of the current flowing through the circuit capacitor.
 5. Thelow-loss current sensor of claim 4, wherein the peak detector includes:a diode having an input connected to the differentiator output; and adetection capacitor connected between the diode and ground, the voltageacross the detection capacitor representing a peak detected currentvalue.
 6. The low-loss current sensor of claim 4, wherein the peakdetector includes a switch connected to the detection capacitor fordischarging the detection capacitor prior to sensing a new currentflowing through the differentiator circuit.
 7. The low-loss currentsensor of claim 4, wherein the peak detector includes a switch connectedacross the detection capacitor, further comprising a controller adaptedto sample a voltage across the detection capacitor and to close theswitch to discharge the detection capacitor prior to sampling thevoltage.
 8. The low-loss current sensor of claim 1, further comprising acontroller having an input connected to the differentiator circuitoutput and adapted to generate a digital value corresponding to thecurrent flowing through the circuit capacitor.
 9. The low-loss currentsensor of claim 8, wherein: the circuit having the capacitor includes azero-voltage-switching power converter having first and second switchesconnected in series and defining a common node therebetween, the commonnode being connected to the input of the differentiator circuit; thecontroller is adapted to determine an optimal energy transfer point byvarying the turn-on time of at least one of the first and secondswitches and comparing corresponding sensed current values of thedifferentiator output.
 10. The low-loss current sensor of claim 1,wherein the capacitance of the sensing capacitor is at most one percentof the capacitance of the circuit capacitor.
 11. A power convertercomprising: first and second switches connected in series and defining acommon node therebetween; at least one circuit capacitor connected toone of the first and second switches; an inductor connected to thecommon node and adapted to provide power to a load or receive power froman input; a current sensor including: a sensing capacitor having a firstterminal connected to the common node and adapted to generate an outputcarrying a current which is proportional to the current flowing throughthe first and second capacitors; a resistor connected to the secondterminal of the sensing capacitor and defining an output of the currentsensor, a voltage drop across the resistor being substantiallyproportional to the current flowing through the sensing capacitor; acontroller adapted to switch the first and second switches in acomplementary manner and control the on time of at least one of thefirst and second switches based on readings of the current sensoroutput.
 12. The power converter of claim 11, wherein the controller isadapted to determine an optimal energy transfer point by varying theturn-on time of at least one of the first and second switches andcomparing corresponding values of the current sensor output.
 13. Thepower converter of claim 11, further comprising a peak detector coupledto the resistor to detect a negative or positive peak value of thecurrent flowing through the at least one circuit capacitor.
 14. Thepower converter of claim 13, wherein the peak detector includes: a diodehaving an input connected to the resistor; and a detection capacitorconnected between the diode and a signal reference, the voltage acrossthe detection capacitor representing a peak detected current value. 15.The power converter of claim 14, wherein the peak detector includes aswitch connected to the detection capacitor for discharging thedetection capacitor prior to sensing a new current flowing through thesensing capacitor.
 16. The power converter of claim 11, wherein thecapacitance of the sensing capacitor is at most one percent of thecapacitance of the at least one circuit capacitor.
 17. The powerconverter of claim 11, wherein the controller is adapted to dynamicallyand continuously vary a switching frequency of the first and secondswitches based on readings of the current sensor output.
 18. The powerconverter of claim 11, wherein the controller is adapted to sample thecurrent sensor output only during dead times of the first and secondswitches.
 19. The power converter of claim 11, wherein the controllerconverts the current sensor output to digital values and samples thesampled digital values.
 20. The power converter of claim 14, wherein thecontroller varies the turn-on time of at least one of the first andsecond switches based on a direct comparison of a present value of thevoltage across the detection capacitor and a previous value of thevoltage across the detection capacitor without further calculation.